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MIC26400
5A Hyper Speed ControlTM Synchronous DC/DC Buck Regulator SuperSwitcher IITM
General Description
The Micrel MIC26400 is a constant-frequency, synchronous buck regulator featuring a unique digitally modified adaptive ON-time control architecture. The MIC26400 operates over an input supply range of 4.5V to 26V and provides a regulated output at up to 5A of output current. The output voltage is adjustable down to 0.8V with a typical accuracy of 1%, and the device operates at a switching frequency of 300kHz. Micrel's Hyper Speed ControlTM architecture allows for ultrafast transient response while reducing the output capacitance and also makes (High VIN)/(Low VOUT) operation possible. This digitally modified adaptive tON ripple control architecture combines the advantages of fixed frequency operation and fast transient response in a single device. The MIC26400 offers a full suite of protection features to ensure protection of the IC during fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, foldback current limit, "hiccup" mode shortcircuit protection and thermal shutdown. All support documentation can be found on Micrel's web site at: www.micrel.com.
Features
* Hyper Speed ControlTM architecture enables - High delta V operation (VIN = 26V and VOUT = 0.8V) - Small output capacitance 4.5V to 26V input voltage Output down to 0.8V with 1% accuracy Any CapacitorTM Stable - Zero ESR to high-ESR output capacitance 5A output current capability 300kHz switching frequency Internal compensation Up to 95% efficiency 6ms Internal soft-start Foldback current limit and "hiccup" mode short-circuit protection Thermal shutdown Supports safe start-up into a pre-biased load -40C to +125C junction temperature range 28-pin 5mm X 6mm MLF(R) package
* * * * * * * * * * * * *
Applications
* Distributed power systems * Communications/networking infrastructure * Set-top box, gateways and routers * Printers, scanners, graphic cards and video cards ____________________________________________________________________________________________________________
Typical Application
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc. MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc. Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
July 2010
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Micrel, Inc.
MIC26400
Ordering Information
Part Number MIC26400YJL Voltage Adjustable Switching Frequency 300kHz Junction Temperature Range -40C to +125C Package 28-Pin 5mm X 6mm MLF(R) Lead Finish Pb-Free
Pin Configuration
28-Pin 5mm X 6mm MLF(R) (YJL)
Pin Description
Pin Number 13, 14, 15, 16, 17, 18, 19 24 Pin Name PVIN Pin Function High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from 4.5V to 26V. Input capacitors between the PVIN pins and the power ground (PGND) are required. Note that the connection must be kept short. Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is reduced (typically 0.7mA). Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer. (see PCB Layout Guidelines for details.) VDD Bias (Input): Power to the internal reference and control sections of the MIC26400. The VDD operating voltage range is from 4.5V to 5.5V. A 2.2F ceramic capacitor from the VDD pin-to-PGND must be placed next to the IC. Power Ground. PGND is the ground path for the MIC26400 buck converter power stage. The PGND pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (SGND) loop. Current Sense (Input): High current output driver return. The CS pin connects directly to the switch node. Due to the high speed switching on this pin, the CS pin should be routed away from sensitive nodes. CS pin also senses the current by monitoring the voltage across the low-side internal MOSFET during OFF-time. Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1F is connected between the BST pin and the SW pin. Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET drain. Power Supply Voltage (Input): Requires bypass capacitor to SGND. No Connect.
EN
25 26 27
FB SGND VDD
2, 5, 6, 7, 8, 21
PGND
22
CS
20 4, 9, 10, 11, 12 23 1, 3, 28
BST SW VIN NC
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Absolute Maximum Ratings(1, 2)
PVIN to PGND................................................ -0.3V to +28V VIN to PGND ....................................................-0.3V to PVIN VDD to PGND ................................................... -0.3V to +6V VSW, VCS to PGND .............................. -0.3V to (PVIN +0.3V) VBST to VSW ........................................................ -0.3V to 6V VBST to PGND .................................................. -0.3V to 34V VEN to PGND ...................................... -0.3V to (VDD + 0.3V) VFB to PGND....................................... -0.3V to (VDD + 0.3V) PGND to SGND ........................................... -0.3V to +0.3V Junction Temperature .............................................. +150C Storage Temperature (TS).........................-65C to +150C Lead Temperature (soldering, 10sec)........................ 260C
Operating Ratings(3)
Supply Voltage (PVIN, VIN)................................. 4.5V to 26V Bias Voltage (VDD)............................................ 4.5V to 5.5V Enable Input (VEN) ................................................. 0V to VDD Junction Temperature (TJ) ........................ -40C to +125C Maximum Power Dissipation......................................Note 4 Package Thermal Resistance(4) 5mm x 6mm MLF(R) (JA) ....................................36C/W
Electrical Characteristics(5)
PVIN = VIN =12V, VDD = 5V; VBST - VSW = 5V; TA = 25C, unless noted. Bold values indicate -40C TJ +125C.
Parameter Power Supply Input Input Voltage Range (VIN, PVIN) VDD Bias Voltage Operating Bias Voltage (VDD) Under-Voltage Lockout Trip Level UVLO Hysteresis Quiescent Supply Current Shutdown Supply Current Reference Feedback Reference Voltage Load Regulation Line Regulation FB Bias Current Enable Control EN Logic Level High EN Logic Level Low EN Bias Current Oscillator Switching Frequency Maximum duty cycle Minimum duty cycle Minimum Off-time Soft-Start Soft-Start time 6 ms
(6) (7)
Condition
Min.
Typ.
Max.
Units
4.5 4.5 VDD Rising VFB = 1.5V VDD = VBST = 5.5V, VIN = 26V SW = unconnected, VEN = 0V 0C TJ 85C (1.0%) -40C TJ 125C (1.5%) IOUT = 0A to 5A VIN = (VOUT + 3.0V) to 26V VFB = 0.8V 4.5V < VDD < 5.5V 4.5V < VDD < 5.5V VEN = 0V 225 VFB = 0V VFB > 0.8V 1.2 0.792 0.788 2.4 5 2.7 50 1.4 0.7
26 5.5 3.2 3 2
V V V mV mA mA
0.8 0.8 0.2 0.1 5 0.85 0.78 50 300 87 0 360
0.808 0.812
V % % nA V
0.4
V A
375
kHz % % ns
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Electrical Characteristics(5) (Continued)
PVIN = VIN =12V, VDD = 5V; VBST - VSW = 5V; TA = 25C, unless noted. Bold values indicate -40C TJ +125C.
Parameter Short Circuit Protection Current-Limit Threshold Short Circuit Current Internal FETs Top-MOSFET RDS (ON) Bottom-MOSFET RDS (ON) SW Leakage Current VIN Leakage Current Thermal Protection Over-temperature Shutdown Over-temperature Shutdown Hysteresis
Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 3. The device is not guaranteed to function outside operating range. 4. PD(MAX) = (TJ(MAX) - TA)/ JA, where JA depends upon the printed circuit layout. See "Applications Information." 5. Specification for packaged product only. 6. Measured in test mode. 7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
Condition
Min.
Typ.
Max.
Units
VFB = 0.8V VFB = 0V ISW = 1A ISW = 1A VIN = 26V, VSW = 26V, VEN = 0V, VBST = 31.5 V VIN = 26V, VSW = 0V, VEN = 0V, VBST = 31.5V TJ Rising
6
13 6 43 12.5 60 25 155 10
A A m m A A C C
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Typical Characteristics
VIN Operating Supply Current vs. Input Voltage
10.0 SHUTDOWN CURRENT (A) SUPPLY CURRENT (mA) 8.0 6.0 4.0
VOUT = 1.2V
VIN Shutdown Current vs. Input Voltage
20 16 12 8 4 0
VDD = 5V VEN = 0V
VDD Operating Supply Current vs. Input Voltage
10 SUPPLY CURRENT (mA) 8 6 4
VOUT = 1.2V
2.0 0.0 4 10 16
IOUT = 0A VDD = 5V SWITCHING
2 0
VDD = 5V SWITCHING
22
28
4
10
16
22
28
4
10
16
22
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage vs. Input Voltage
0.808 FEEDBACK VOLTAGE (V) TOTAL REGULATION (%) 1.0% 0.8% 0.6% 0.4% 0.2% 0.0% 4 10 16 22 28 4
Total Regulation vs. Input Voltage
20
VOUT = 1.2V
Current Limit vs. Input Voltage
0.804
CURRENT LIMIT (A)
VDD = 5V IOUT = 0A to 5A
15
0.800
10
0.796
VOUT = 1.2V VDD = 5V IOUT = 0A
5
VOUT = 1.2V VDD = 5V
0.792 INPUT VOLTAGE (V)
0 10 16 22 28 4 10 16 22 28 INPUT VOLTAGE (V) INPUT VOLTAGE (V)
Switching Frequency vs. Input Voltage
390
SWITCHING FREQUENCY (kHz)
VOUT = 1.2V
345
VDD = 5V IOUT = 0A
300
255
210 4 10 16 22 28 INPUT VOLTAGE (V)
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Typical Characteristics (Continued)
VDD Operating Supply Current vs. Temperature
10.0 SUPPLY CURRENT (mA)
VIN = 12V VDD = 5V IOUT = 0A SWITCHING
VDD Shutdown Current vs. Temperature
1 2.8 2.7 2.6 2.5 2.4 2.3 -50 -20 10 40 70 100 130 -50 SUPPLY CURRENT (mA)
VDD UVLO Threshold vs. Temperature
6.0 4.0 2.0 0.0 -50 -20 10 40
0.6 0.4 0.2 0
VIN = 12V IOUT = 0A VDD = 5V VEN = 0V
VDD THRESHOLD (V)
8.0
VOUT = 1.2V
0.8
Rising
Falling
VIN = 12V
70
100
130
-20
10
40
70
100
130
TEMPERATURE (C)
TEMPERATURE (C)
TEMPERATURE (C)
VIN Operating Supply Current vs. Temperature
10.0 SUPPLY CURRENT (mA) 8.0 6.0 4.0 2.0 0.0 -50 -20 10 40 70 100 130 TEMPERATURE (C)
VOUT = 1.2V VDD = 5V IOUT = 0A SWITCHING
VIN Shutdown Current vs. Temperature
10.0 SUPPLY CURRENT (A) 25 20 15 10
Current Limit vs. Temperature
VIN = 12V
6.0 4.0
VIN = 12V
CURRENT LIMIT (A)
8.0
VIN = 12V
2.0 0.0 -50 -20 10 40
VDD = 5V IOUT = 0A
5 0
VOUT = 1.2V VDD = 5V
70
100
130
-50
-20
10
40
70
100
130
TEMPERATURE (C)
TEMPERATURE (C)
Feedback Voltage vs. Temperature
0.808 FEEDBACK VOLTAGE (V) LOAD REGULATION (%)
VIN = 12V
Load Regulation vs. Temperature
1.0% 0.8% 0.6% 0.4% 0.2% 0.0% LINE REGULATION (%)
VIN = 12V VOUT = 1.2V VDD = 5V IOUT = 0A to 5A
Line Regulation vs. Temperature
0.5% 0.4% 0.3% 0.2% 0.1% 0.0%
VIN = 6V to 26V VOUT = 1.2V VDD = 5V
0.804
VOUT = 1.2V VDD = 5V IOUT = 0A
0.800
0.796
0.792 -50 -20 10 40 70 100 130 TEMPERATURE (C)
-50
-20
10
40
70
100
130
-50
-20
10
40
70
100
130
TEMPERATURE (C)
TEMPERATURE (C)
Switching Frequency vs. Temperature
345
SWITCHING FREQUENCY (kHz)
EN Bias Current vs. Temperature
100
330 315 300 285 270 255 -50 -20 10 40
V OUT = 1.2V V DD = 5V IOUT = 0A
EN BIAS CURRENT (A)
V IN = 12V
80 60 40
VIN = 12V
20 0
VOUT = 1.2V VDD = 5V
70
100
130
-50
-20
10
40
70
100
130
TEMPERATURE (C)
TEMPERATURE (C)
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Typical Characteristics (Continued)
Efficiency vs. Output Current
100
Feedback Voltage vs. Output Current
0.808 FEEDBACK VOLTAGE (V)
OUTPUT VOLTAGE (V) 1.212 1.208 1.204 1.2 1.196 1.192 1.188
Output Voltage vs. Output Current
95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 0 1 2 3 4 5 OUTPUT CURRENT (A)
VOUT = 1.2V VDD = 5V 12VIN 24VIN
0.804
0.800
0.796
VIN = 12V VOUT = 1.2V VDD = 5V
VIN = 12V VOUT = 1.2V VDD = 5V
0.792 0 1 2 3 4 5
0
1
2
3
4
5
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Line Regulation vs. Output Current
0.5% 0.4% 0.3% 0.2% 0.1% 0.0% 0 1 2 3 4 5 OUTPUT CURRENT (A)
VOUT = 1.2V VDD = 5V
Switching Frequency vs. Output Current
390
SWITCHING FREQUENCY (kHz)
5 4.8 OUTPUT VOLTAGE (V) 4.6 4.4 4.2 4 3.8 3.6 3.4 0 1 2 3 4 5 0 1
Output Voltage (VIN = 5V) vs. Output Current
VIN = 5V VFB < 0.8V VDD = 5V
LINE REGULATION (%)
VIN = 6V to 26V
360 330 300 270 240 210 OUTPUT CURRENT (A)
VIN = 12V VOUT = 1.2V VDD = 5V
TA 25C 85C 125C
2
3
4
5
6
7
OUTPUT CURRENT (A)
Die Temperature* (VIN = 5V) vs. Output Current
60 DIE TEMPERATURE (C) DIE TEMPERATURE (C) 60
Die Temperature* (VIN = 12V) vs. Output Current
60 DIE TEMPERATURE (C)
Die Temperature* (VIN = 24V) vs. Output Current
40
40
40
20
V IN = 5V V OUT = 1.2V V DD = 5V
20
VIN = 12V VOUT = 1.2V VDD = 5V
20
V IN = 24V V OUT = 1.2V V DD = 5V
0 0 1 2 3 4 5 OUTPUT CURRENT (A)
0 0 1 2 3 4 5 OUTPUT CURRENT (A)
0 0 1 2 3 4 5 OUTPUT CURRENT (A)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26400 case mounted on a 5 square inch PCB, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components.
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Typical Characteristics (Continued)
Efficiency (VIN = 5V) vs. Output Current
100 100 95 EFFICIENCY (%) EFFICIENCY (%) 95
3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V
Efficiency (VIN = 12V) vs. Output Current
5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V
Efficiency (VIN = 24V) vs. Output Current
95 90 EFFICIENCY (%) 85 80 75 70
5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V
90 85 80 75 70
90
85
80 0 1 2 3 4 5 6 7 OUTPUT CURRENT (A)
0
1
2
3
4
5
6
7
0
1
2
3
4
5
6
7
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
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Functional Characteristics
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Functional Characteristics (Continued)
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Functional Characteristics (Continued)
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Functional Diagram
Figure 1. MIC26400 Block Diagram
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MIC26400 The maximum duty cycle is obtained from the 360ns tOFF(min):
D max = t S - t OFF(min) tS = 1- 360ns tS
Functional Description
The MIC26400 is an adaptive ON-time synchronous step-down DC/DC regulator. It is designed to operate over a wide input voltage range from 4.5V to 26V and provides a regulated output voltage at up to 5A of output current. A digitally modified adaptive ON-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. Over-current protection is implemented without the use of an external sense resistor. The device includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. Theory of Operation Figure 1 illustrates the block diagram for the control loop of the MIC26400. The output voltage is sensed by the MIC26400 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low gain transconductance (gm) amplifier. If the feedback voltage decreases and the output of the gm amplifier is below 0.8V, then the error comparator will trigger the control logic and generate an ON-time period. The ON-time period length is predetermined by the "FIXED tON ESTIMATION" circuitry:
t ON(estimated) = VOUT VIN x 300kHz
(2)
(1)
Where VOUT is the output voltage and VIN is the power stage input voltage. At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 360ns, the MIC26400 control logic will apply the tOFF(min) instead. tOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET.
where tS = 1/300kHz = 3.33s. It is not recommended to use MIC26400 with a OFF-time close to tOFF(min) during steady-state operation. The actual ON-time and resulting switching frequency will vary with the part-to-part variation in the rise and fall times of the internal MOSFETs, the output load current, and variations in the VDD voltage. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications, such as 26V to 1.0V. The minimum tON measured on the MIC26400 evaluation board is about 184ns. During load transients, the switching frequency is changed due to the varying OFF-time. To illustrate the control loop operation, we will analyze both the steady-state and load transient scenarios. For easy analysis, the gain of the gm amplifier is assumed to be 1. With this assumption, the inverting input of the error comparator is the same as the feedback voltage. Figure 2 shows the MIC26400 control loop timing during steady-state operation. During steady-state, the gm amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry.
Figure 2. MIC26400 Control Loop Timing
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MIC26400
Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. The MIC26400 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with 9.7mV steps. Therefore, the output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VDD must be powered up at the same time or after VIN to make the soft-start function correctly. Current Limit The MIC26400 uses the RDS(ON) of the internal low-side power MOSFET to sense over-current conditions. This method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. The low-side MOSFET is used because it displays much lower parasitic oscillations during switching than the high-side MOSFET. In each switching cycle of the MIC26400 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. If the inductor current is greater than 13A, then the MIC26400 turns off the highside MOSFET and a soft-start sequence is triggered. This mode of operation is called "hiccup mode" and its purpose is to protect the downstream load in case of a hard short. The current-limit threshold has a foldback characteristic related to the feedback voltage, as shown in Figure 4.
Figure 3. MIC26400 Load Transient Response
Unlike true current-mode control, the MIC26400 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor current ripple if the ESR of the output capacitor is large enough. The MIC26400 control loop has the advantage of eliminating the need for slope compensation. In order to meet the stability requirements, the MIC26400 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended feedback voltage ripple is 20mV~100mV. If a low-ESR output capacitor is selected, the feedback voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. In these cases, ripple injection is required to ensure proper operation. Please refer to "Ripple Injection" subsection in "Application Information" for more details about the ripple injection technique.
Current-Limit Thresold vs. Feedback Voltage
20.0 CURRENT-LIMIT THRESHOLD (A)
16.0
12.0
8.0
4.0
0.0 0.0 0.2 0.4 0.6 0.8 1.0 FEEDBACK VOLTAGE (V)
Figure 4. MIC26400 Current Limit Foldback Characteristic
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MIC26400 the power stroke (high-side switching) cycle, i.e. BST = 10mA x 3.33s/0.1F = 333mV. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the VDD supply voltage. The nominal low-side gate drive voltage is VDD and the nominal high-side gate drive voltage is approximately VDD - VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs.
MOSFET Gate Drive The Block Diagram of Figure 1 shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode D1 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1F to 1F is sufficient to hold the gate voltage with minimal droop for
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MIC26400 but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below:
2 PINDUCTOR(Cu) = IL(RMS) x RWINDING
Application Information
Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below:
VOUT x (VIN(max) - VOUT ) VIN(max) x fsw x 20% x IOUT(max)
(8)
The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. PWINDING(Ht) = RWINDING(20C) x (1 + 0.0042 x (TH - T20C))
L=
(4)
(9)
Where: fSW = switching frequency, 300kHz 20% = ratio of AC ripple current to DC output current VIN(max) = maximum power stage input voltage The peak-to-peak inductor current ripple is:
VOUT x (VIN(max) - VOUT ) VIN(max) x fsw x L
Where: TH = temperature of wire under full load T20C = ambient temperature RWINDING(20C) = room temperature winding resistance (usually specified by the manufacturer) (5) Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor's ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated:
VOUT(pp) IL(PP)
IL(pp) =
The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(pk) =IOUT(max) + 0.5 x IL(pp) (6)
The RMS inductor current is used to calculate the I2R losses in the inductor.
IL(PP) 12
2
ESR COUT IL(RMS) = IOUT(max) +
2
(10)
(7) Where: VOUT(pp) = peak-to-peak output voltage ripple IL(PP) = peak-to-peak inductor current ripple
Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC26400 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used July 2010 16
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Micrel, Inc. The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated below:
IL(PP) 2 + IL(PP) x ESR C = OUT C x f SW x 8 OUT (11)
2
MIC26400
The peak input current is equal to the peak inductor current, so: VIN = IL(pk) x ESRCIN (14)
VOUT(pp)
(
)
Where: D = duty cycle COUT = output capacitance value fSW = switching frequency As described in the "Theory of Operation" subsection in "Functional Description", the MIC26400 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Please refer to the "Ripple Injection" subsection for more details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated below:
The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low:
ICIN(RMS) IOUT(max) x D x (1 - D)
(15)
The power dissipated in the input capacitor is: PDISS(CIN) = ICIN(RMS)2 x ESRCIN (16)
Voltage Setting Components The MIC26400 requires two resistors to set the output voltage as shown in Figure 5.
ICOUT (RMS) =
IL(PP) 12
Figure 5. Voltage-Divider Configuration
(12)
The output voltage is determined by the equation:
VOUT = VFB x (1 + R1 ) R2
The power dissipated in the output capacitor is:
PDISS(COUT ) = ICOUT (RMS) x ESR COUT
2
(17)
(13)
Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor's voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor's ESR. July 2010 17
Where VFB = 0.8V. A typical value of R1 can be between 3k and 10k. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using:
R2 = VFB x R1 VOUT - VFB
(18)
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MIC26400
Ripple Injection The VFB ripple required for proper operation of the MIC26400 gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as a 1V, the output voltage ripple is only 10mV to 20mV, and the feedback voltage ripple is less than 20mV. If the feedback voltage ripple is so small that the gm amplifier and error comparator can't sense it, the MIC26400 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: 1) Enough ripple at the feedback voltage due to the large ESR of the output capacitors. As shown in Figure 6a, the converter is stable without any ripple injection. The feedback voltage ripple is:
VFB(pp) = R2 x ESR COUT x IL (pp) R1 + R2
Figure 6b. Inadequate Ripple at FB
(19)
Figure 6c. Invisible Ripple at FB
where IL(pp) is the peak-to-peak value of the inductor current ripple. 2) Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feedforward capacitor Cff in this situation, as shown in Figure 6b. The typical Cff value is between 1nF and 100nF. With the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple:
VFB(pp) ESR x IL (pp)
In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 6c. The injected ripple is:
VFB(pp) = VIN x K div x D x (1 - D) x 1 fSW x
(21)
K div =
(20)
R1//R2 R inj + R1//R2
(22)
3) Virtually no ripple at the FB pin voltage due to the very low ESR of the output capacitors.
Where: VIN = Power stage input voltage D = duty cycle fSW = switching frequency = (R1//R2//Rinj) x Cff In equations (21) and (22), it is assumed that the time constant associated with Cff must be much greater than the switching period:
1 T = << 1 fSW x
(23)
Figure 6a. Enough Ripple at FB
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Micrel, Inc. If the voltage divider resistors R1 and R2 are in the k range, a Cff of 1nF to 100nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in k range. Step 2. Select Rinj according to the expected feedback voltage ripple using equation (24),
VFB(pp) VIN fSW x D x (1 - D)
MIC26400
Thermal Measurements Measuring the IC's case temperature is recommended to insure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get erroneous results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. Two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire or higher (smaller wire size) to minimize the wire heat-sinking effect. In addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ICs. However, a IR thermometer from Optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time.
K div =
x
(24)
Then the value of Rinj is obtained as:
R inj = (R1//R2) x ( 1 K div - 1)
(25)
Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies.
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Inductor
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC26400 converter. IC
* * * *
Keep the inductor connection to the switch node (SW) short. Do not route any digital lines underneath or close to the inductor. Keep the switch node (SW) away from the feedback (FB) pin. The CS pin should be connected directly to the SW pin to accurate sense the voltage across the lowside MOSFET.
*
The 2.2F ceramic capacitor, which is connected to the VDD pin, must be located right at the IC. The VDD pin is very noise sensitive and placement of the capacitor is very critical. Use wide traces to connect to the VDD and PGND pins. The signal ground pin (SGND) must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer. Place the IC close to the point of load (POL). Use fat traces to route the input and output power lines. Signal and power grounds should be kept separate and connected at only one location. Place the input capacitor next. Place the input capacitors on the same side of the board and as close to the IC as possible. Keep both the PVIN and PGND connections short. Place several vias to the ground plane close to the input capacitor ground terminal. Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. In "Hot-Plug" applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied.
To minimize noise, place a ground plane underneath the inductor. Output Capacitor * Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM.
*
*
*
* * *
Input Capacitor
The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. RC Snubber * Place the RC snubber on the same side of the board and as close to the SW pin as possible.
*
* * * * * *
*
*
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MIC26400
Evaluation Board Schematic
Figure 7. Schematic of MIC26400 Evaluation Board (J13, R13, R15 are for testing purposes)
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MIC26400
Bill of Materials
Item Part Number Manufacturer Description Qty
C1 C2, C3 C4, C5, C13
B41125A7227M 12105C475KAZ2A GRM32ER71H475KA88L 12106D107MAT2A GRM32ER60J107ME20L 06035C104KAT2A GRM188R71H104KA93D C1608X7R1H104K 0805ZC225MAT2A
EPCOS AVX
(1)
220F Aluminum Capacitor, SMD, 35V 4.7F Ceramic Capacitor, X7R, Size 1210, 50V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V
1 2 1
(2)
Murata(3) AVX Murata AVX Murata TDK
(4)
C6, C7, C10
0.1F Ceramic Capacitor, X7R, Size 0603, 50V
4
AVX Murata TDK AVX Murata TDK AVX Murata TDK 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 2.2F Ceramic Capacitor, X7R, Size 0805, 10V 2
C8, C9
GRM21BR71A225KA01L C2012X7R1A225K 06035C102KAT2A
C11
GRM188R71H102KA01D C1608X7R1H102K 06035C223KAZ2A
C12 C14 C15 D1 D2 L1 Q1 R1 R2, R16 R3, R4 R5 R6 R7 R8
GRM188R71H223K C1608X7R1H223K Open Open SD103AWS-7 SD103AWS CMDZ5L6 HCF1305-4R0-R FCX619 CRCW06034R75FKEA CRCW08051R21FKEA CRCW060310K0FKEA CRCW060380K6FKEA CRCW060340K2FKEA CRCW060320K0FKEA CRCW060311K5FKEA
Diodes Inc(6) Vishay(7) Central Semi(8) ZETEX Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale
Small Signal Schottky Diode 5.6V Zener Diode 50V NPN Transistor 4.75 Resistor, Size 0603, 1% 1.21 Resistor, Size 0805, 1% 10k Resistor, Size 0603, 1% 80.6k Resistor, Size 0603, 1% 40.2k Resistor, Size 0603, 1% 20k Resistor, Size 0603, 1% 11.5k Resistor, Size 0603, 1%
1 1 1 1 1 1 2 1 1 1 1
Cooper Bussmann(9) 4.0H Inductor, 12A Saturation Current
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Bill of Materials (Continued)
Item Part Number Manufacturer Description Qty
R9 R10 R11 R12 R13 R14 R15
U1
Notes:
CRCW06038K06FKEA CRCW06034K75FKEA CRCW06033K24FKEA CRCW06031K91FKEA CRCW06030000FKEA CRCW06035K23FKEA CRCW060349R9FKEA
MIC26400YJL
Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale
Micrel. Inc.
(10)
8.06k Resistor, Size 0603, 1% 4.75k Resistor, Size 0603, 1% 3.24k Resistor, Size 0603, 1% 1.91k Resistor, Size 0603, 1% 0 Resistor, Size 0603, 5% 5.23k Resistor, Size 0603, 1% 49.9 Resistor, Size 0603, 1%
26V/5A Synchronous Buck DC/DC Regulator
1 1 1 1 1 1 1
1
1. EPCOS: www.epcos.com. 2. AVX: www.avx.com. 3. Murata: www.murata.com. 4. TDK: www.tdk.com. 5. SANYO: www.sanyo.com. 6. Diode Inc.: www.diodes.com. 7. Vishay: www.vishay.com. 8. Central Semi: www.centralsemi.com. 9. Cooper Bussmann: www.cooperbussmann.com. 10. Micrel, Inc.: www.micrel.com.
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PCB Layout
Figure 8. MIC26400 Evaluation Board Top Layer
Figure 9. MIC26400 Evaluation Board Mid-Layer 1 (Ground Plane)
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MIC26400
PCB Layout (Continued)
Figure 10. MIC26400 Evaluation Board Mid-Layer 2
Figure 11. MIC26400 Evaluation Board Bottom Layer
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MIC26400
Recommended Land Pattern
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Package Information
28-Lead 5mm x 6mm MLF(R) (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2010 Micrel, Incorporated.
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